1. Field of the Invention
The present invention relates to a digital demodulator of a wireless mobile system of a code division multiple connection system and, in more particularly to, a .pi./4 quadrature phase shift keying (QPSK) digital demodulation apparatus and a method thereof, in which error data of modulated signals may be stably corrected and recovered by a .pi./4 QPSK without any additional error correction circuits.
2. Description of the Conventional Art
Recently, there is a tendency that the necessity of mobile communication system service is increasing rapidly. Therefore, in order to realize the mobile communication service to a plurality of subscribers, a modulation system having high spectrum efficiency is required so that it becomes essential to develop techniques to utilize frequency efficiency such as the spectrum efficiency and error correction encoding/decoding by using high efficiency of digital modulation/demodulation.
In order to realize the mobile communication service worldwide, group special groups are decided by digital cellular standards in Europe and Telecommunication Industry Association (TIA) is adopted as a standard of a land mobile system in the United States.
As the digital mobile modulation/demodulation method, Gaussian-prefiltered Minimum Shift Keying method of Europe and .pi./4 shift QPSK method of the United States and Japan are typical.
Generally, smaller bandwidths are transmitted by a linear modulation transmit rather than by a constant envelope modulation such as the quadrature amplitude shift keying (QMSK) and Tamed Frequency Modulation (TFM). However, the constant envelope modulation may adopt a nonlinear amplifier which has higher power efficiency even though its precision for amplitudes is not good since modulated signals of the constant envelope modulation include any information in their amplitudes.
On the other hand, the linear modulation adopts a linear amplifier which is good for keeping signal waveforms precisely while modulating signal amplitudes. Under the circumstances, .pi./4 QPSK shift QPSK has been proposed for the digital cellular mobile communications.
In general, the digital communication system is subject to coherent detection in order to use bandwidth and the power efficiently.
Even though synchronization system has good power efficiency for white Gaussian noise theoretically, synchronization efficiency decreases when phase noise exists in case of multipath fading or doppler shift, especially in a narrowband mobile communication system.
An asynchronous system of differential detection is properly used for the narrowband mobile communication system since carrier recovery is not necessary in this system. On the other hand, the linear modulation such as the QPSK has better bandwidth efficiency comparing with the constant envelope modulation such as the QMSK or the TEM.
Even though the digital communication system adopts the nonlinear amplifier for its power efficiency, if a linear modulated carrier is passed through this nonlinear amplifier to increase the power efficiency, it has disadvantages that spectrum is spread and cophase components and orthogonal components are degraded so that serious interface is caused between adjacent channels. Therefore, it becomes impossible to use the spectrum and bit error rate (BER) characteristics decrease due to the nonlinear amplifier.
In order to solve these problems of the nonlinear modulation, offset QPSK (OQPSK) has been proposed to reduce the change of carrier. However, the OQPSK modulation method requires the coherent detection which has low bit error rate (BER) characteristics in the mobile communication channels.
The constant envelope modulation such as the minimum shift keying (MSK), QMSK or TEM may use the nonlinear amplifier without the spectrum dispersion and the differential detection but has still problem that the spectrum efficiency is low.
Therefore, in order to resolve the above disadvantages, .pi./4 QPSK shift QPSK has been proposed as a linear modulation method which may utilize the nonlinear amplifier and keep the power efficiency and bandwidth efficiency high.
The .pi./4 QPSK shift QPSK modulation method is limited to .+-..pi./4 and .+-.3 .pi./4 in the phase change and not subject to 55 change so that the amplitude change decreases. Further, this .pi./4 QPSK shift QPSK modulation method may use both the coherent detection and differential detection so that the bandwidth efficiency is very excellent.
Now, the conventional .pi./4 QPSK shift QPSK modulation apparatus is explained with reference to FIG. 1. In FIG. 1, the .pi./4 QPSK shift QPSK modulation apparatus includes a serial/parallel conversion part 101 to separate data serial input data from an input terminal 100 into orthogonal component data SQ and cophase component data SI to output them parallel, an automatic differential encoding part 102 to obtain a phase shift keying amount .DELTA.QK from the orthogonal component data and the cophase component data which are separately input from the serial/parallel conversion part 101, signal mapping part 103 to output a baseband signal Q(t) of orthogonal component and a baseband signal I(t) of cophase component for time on the basis of the phase shift keying amount and an immediately previous signal, a first low-pass filter part 104 and a second low-pass filter part 105 for respectively passing through the baseband signal Q(t) of orthogonal component and the baseband signal I(t) of cophase component, a first conversion part 107 and a second conversion part 109 to modulate with high frequency signals the baseband signal Q(t) of orthogonal component and the baseband signal I(t) of cophase component which are respectively filtered through the low-pass filter part 104 and 105 and input from input terminals 106 and 108, and a synthesis part 110 to synthesize the modulated output signal from the first and second conversion parts 107 and 109 for providing them to an antenna ANT.
In the conventional modulation apparatus as described above, input serial data SI from the input terminal 100 are divided into the cophase component data SQ and the orthogonal component data SQ through the serial/parallel conversion part 101 and provided to the differential encoding part 102.
The differential encoding part 102 encodes the cophase component data SQ and the orthogonal component data SQ to obtain a phase shift keying amount .DELTA.QK from the phase of the immediately previous signal by Gray Codes and provides the obtained phase shift keying amount .DELTA.QK to the signal mapping part 103.
The signal mapping part 103 operates the phase shift keying amount a .DELTA.QK obtained by the differential encoding part 102 and the immediately previous signal to output unfiltered baseband non-return-to zero signals I(t) and Q(t) which respectively show the cophase component and the orthogonal component.
The baseband signals of the cophase component I(t) and the orthogonal component Q(t) are respectively filtered by the first and the second low-pass filter parts 104 and 105 to be output to the first and the second conversion parts 107 and 109.
Then, the first conversion part 107 mixes the baseband signals of the cophase component I(t) with a cophase signal cos .omega..sub.o t which is input from the input terminal 106 to output it to the synthesization part 110 while the second conversion part 109 mixes the orthogonal component Q(t) with an orthogonal signal -sin .omega..sub.o t which is input from the input terminal 108 to output it to the synthesization part 110.
Therefore, the synthesization part 110 synthesizes the values modulated by the first and the second conversion parts 107 and 109 to transmit the synthesized values through the antenna ANT.
As described hereinabove, the .pi./4 QPSK shift QPSK modulation apparatus transmits information being included in the phase difference of two continuous channel signals. Therefore, a receiver should detect the phase difference in order to derive the information.
Now, a conventional .pi./4 QPSK demodulation apparatus for detecting the phase difference will be described with reference to FIG. 2.
In FIG. 2, the conventional .pi./4 QPSK demodulation apparatus includes a reception filter part 201 to remove unnecessary signals from a QPSK signal which is input from a receiving input terminal 200, a local oscillation part 203 for generating a local carrier signal, a first phase shift keying part 204 to transit a phase of the local oscillated carrier signal input from the local oscillation part 203 by .pi./2 phase, a first mix part 202 to mix the shifted local carrier signal output from the first phase shift keying part 204 with the modulated signal output from the reception filter part 201, a second mix part 205 to mix the local carrier signal output from the local oscillation part 203 with the modulated signal output from the reception filter part 201, a third low-pass filter part 206 to block carrier component from the signal mixed by the first mix part 202 for deriving a baseband signal of orthogonal component, a fourth low-pass filter part 207 to block carrier component from the signal mixed by the second mix part 205 for deriving baseband signal of cophase component I, a carrier recovery part 220 having third and fourth mix parts 210 and 211 and a loop filtering part 209 to control outputs from the local oscillation part 203 on the basis of phase synchronization of a carrier of a transmitter which is received by the receiver, a clock recovery part 208 to recover a symbol rate clock of the transmitter which is input through the reception filter part 201, a second phase shift keying part 212 to transit a phase of the clock recovered by the clock recovery part 208 by .pi./2 phase to provide to the third mix part 210 of the carrier recovery part 220, a third phase shift keying part 215 to transit a phase of the clock recovered by the clock recovery part 208 by .pi. phase, a second decision part 214 to obtain a phase difference between the clock which is recovered by the clock recovery part 208 and the baseband signal of cophase component I which is derived by the fourth low-pass filter part 207, a first decision part 213 to obtain a phase difference between the phase which is transited by the third phase shift keying part 215 and the baseband signal of orthogonal component which is derived by the third low-pass filter part 206, a first adder 216 to add the phase of cophase component and the phase of orthogonal component which are respectively obtained by the first and the second decision parts 213 and 214, and a second adder 217 to add the signal added by the first adder 216 and the clock which is recovered by the clock recovery part 208, wherein the loop filtering part 209, the third mix part 210, and the fourth mix parts 211 compose the single carrier recovery part 220.
In the conventional demodulation apparatus as described above, when the QPSK modulation signal is input to the reception filter part 201 from the receiving input terminal 200, the reception filter part 201 removes unnecessary signals from the input modulated signal to output to the first and the second mix parts 202 and 205.
The first and the second mix parts 202 and 205 multiply the output from the reception filter part 201 with a synchronized local carrier. In more detail, when the local carrier signal which is oscillated by the local oscillation part 203 is input to the second mix part 205 and the local carrier signal of which phase is shifted by .pi./2 by the first phase shift keying part 204 is input to the first mix part 202, the first mix part 202 mixes the phase shifted local carrier signal and the modulated signal which is input from the reception filter part 201 to derive a signal of orthogonal component.
On the other hand, the second mix part 205 mixes the local carrier signal with the input modulated signal to derive a signal of cophase component. Therefore, the derived orthogonal component signal and the cophase component signal are respectively output to the third and the fourth low-pass filter parts 206 and 207.
The third and the fourth low-pass filter parts 206 and 207 cut off the carrier components of the first and the second mix parts 202 and 205 and derive only the orthogonal component and the cophase component, that is, the baseband signals of Q (quadrature) and I (In phase) channels to send them to the below-mentioned fourth mix part 211 and the first and the second decision parts 213 and 214 of the carrier recovery part 220.
Since the symbol rate clock which is used at the transmitter has a certain error, the clock recovery part 208 recovers this error and provides the recovered clock to the second and the third phase shift keying part 212 and 215, and the second decision part 214.
The second phase shift keying part 212 performs .pi./2 phase shifting of the input clock which is recovered by the clock recovery part 208 to output to the third mix part 210, while the third phase shift keying part 215 performs .pi. phase shifting of the input clock which is recovered by the clock recovery part 208 to output to the first decision part 213. Wherein the fourth mix part 211 mixes the baseband signals of the Q and I channels which are respectively derived by the third and the fourth filter parts 206 and 207 to output to the third mix part 210.
Therefore, the third mix part 210 mixes the output from the fourth mix part 211 with the output from the second phase shift keying part 212 to recover the carrier signal. That is, in a phase synchronized receiver, the frequency and the phase of the local carrier being in use at the receiver have to be synchronized with the frequency and the phase of the carrier being in use at the transmitter.
In order to realize the synchronization, the receiver includes the carrier recovery part 220 having the third and the fourth mix parts 210 and 211 and the loop filter part 209 to recover the frequency and phase of the carrier from the input high frequency signal by using the phase-locked loop (PLL).
On the other hand, the first decision part 213 obtains a phase difference between the phase which is shifted by the third phase shift keying part 215 and the baseband signal of Q channel which is derived by the third low-pass filter 206 to output to the first adder 216, while the second decision part 214 obtains a phase difference between the clock which is recovered by the clock recovery part 208 and the baseband signal of I channel which is derived by the fourth low-pass filter 207 to output to the first adder 216.
Therefore, the first adder 216 adds the orthogonal component phase and the cophase component phase which are respectively obtained by the first and the second decision parts 213 and 214, while the second adder 217 adds the signal which is added by the first adder 216 and the clock of the clock recovery part 208 to output a QPSK modulated signal.
However, the conventional .pi./4 QPSK demodulation apparatus has still disadvantages that its construction is much complicated since the clock recovery part must be provided to recover the erroneous symbol rate clock which is used by the transmitter and the clock recovery part essentially requires the phase-locked loop (PLL). The construction of the conventional .pi./4 QPSK demodulation apparatus is further complicated since an error correction circuit using Hamming Code and Bose-Chaudhuri-Hocquenghem Code (BCH code) must be provided additionally to correct the errors of the modulated signals.
Therefore, hardwares for realizing the phase-locked loop circuit and the error correction circuit are much complicated, resulting in the difficulties of mass production as well as of minimization of the products.